Re: [time-nuts] any WWV audio recordings available?

[Scott Burris]

The GC-1000 keeps declaring the decoding to be invalid partway through 
each minute.
Dean's synthetic WWV gets decoded perfectly and the display turns out 
after 3 complete
minutes.  But the 100Hz level in the synthetic sounds high to me.  Since 
your recording
fades in and out (just like I hear it in LA!), I need to figure out if 
the clock is
having trouble pulling the 100Hz tones out, or if the recording just 
fades too much.


I'm injecting the audio at the volume control, so this is past the AGC, 
so there's
no compensation as the audio fades.  Perhaps that's why I can't decode 
your audio.


Gotta pull out the schematic again and study it.

Scott

Majdi S. Abbas wrote:

On Tue, Feb 09, 2010 at 07:33:07PM -0800, Scott Burris wrote:
  

Does anyone know if there are any > 5min recordings of WWV audio available?
I'm trying to track down some problems with my Heathkit GC-1000 clock and
it sure would be nice if I could inject some known good audio
recording and see if
the clock picks up the time from that.

I've found a couple recordings by googling, but they are both too
short to be useful.



Try this:

(Rather noisy; WWV-5 just now; WWVH-5 in the background.)

http://puck.nether.net/~majdi/WWV-5MHZ.wav

--msa

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Re: [time-nuts] Loran C sounds

[Bill Hawkins]

Two things:

1. The Solar Dynamics Observatory launched this morning, on a mission to
examine the fields and mass ejections that threaten satellites and
power grids. They are delighted to be putting instruments in place
after a prolonged quiet period so they can watch activity ramp up.

The people who shut down non-satellite methods won't be in office at
peak activity.

2. Its all about $$ because they are so easy to count. Decisions based
on dollars don't require any technical background. People with empathy
lose out to people who are focused on the dollars. (You folks with
other currencies can substitute them for declining dollars. The principle
is the same.) Should I automate or retain jobs? Let's see, which costs less
per year? In dollars, that is, not human misery.

Albert J. Bernstein wrote "Dinosaur Brains: Dealing with All Those
Impossible
People at Work" to explain how the most primitive brain of the three in our
skulls sometimes / often gets the upper hand. Fatally attracted to female
fatty deposits? Blame your dino brain and try to rein it in. Trying to deal
with somebody shouting at you? You're watching a dino brain take over. Don't
try to explain, the dino can't hear you unless what you say sounds
threatening.

Dealing with someone who is playing dominance games? Say hello to an ancient
lizard and forget any attempt to reason with the person. Who plays dominance
games? Almost any politician you care to name.

So, no, don't think anything good is gonna happen. There's a reason why
engineers are familiar with not having resources to do it right the first
time.

Darn, the soap box broke.

Bill Hawkins


-Original Message-
From: paul swed
Sent: Thursday, February 11, 2010 9:07 PM

Sure they might take a check for $36M or so.
As they have said without the yanks, its broke. So they will shut down
before Oct.
Its all about $$


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Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[Bob Camp]

Hi

I suspect your noise spike can be cured by a series R-C to ground from the 
junction of Q1 base, Q7 base and all the other stuff. Something is going to 
have to set a high frequency roll off. With no coils some combo of R and C is 
going to have to do it. 

You might also try returning all of the upper emitter resistor bypasses to 
ground rather than B+. Another alternative would be emitter to emitter bypass 
as shown on the JPL schematic. I'm guessing both would improve isolation in a 
real world circuit. 

Bob


On Feb 11, 2010, at 8:34 PM, Bruce Griffiths wrote:

> life speed wrote:
>> Message: 2
>> Date: Fri, 12 Feb 2010 12:12:29 +1300
>> From: Bruce Griffiths
>> The output (collectors of Q5, Q6 emitter of Q4) of the input amplifier
>> sets the dc voltage at the inputs ( Q1 base, Q7 base respectively) of
>> the output amplifiers.
>> 
>> The circuit consists of a unity gain input amplifier (Q4, Q5, Q6) that
>> drives a pair of output amplifiers (Q1, Q2, Q3 and Q7, Q8, Q9
>> respectively) each with a gain of 2x (6dB).
>> The input amplifier is essentially a white emitter follower with a
>> complementary symmetry output stage (shown in transistor electronics
>> books from the 1960's) where an input CE transistor drives a
>> complementary pair of CE transistors with feedback from the common
>> collectors of the 2 output transistors to the input transistor emitter.
>> In effect its merely a very simple unity gain opamp. Its usually best to
>> ensure that the CE output stage pair provide the dominant open loop
>> pole. Using a higher ft (2 to 3x)  input transistor than the output pair
>> is the usual way of ensuring this.
>> 
>> Well, it is so obvious now that you explained it.  I had forgot about the 
>> need for one of the stages to set the dominant pole.
>> 
>> Thanks Bruce and Bob for sharing your obsession with frequency controls.  
>> I'll simulate this further, and have a prototype PCB built within the next 
>> few weeks.  I did notice the resistor at the base of Q2,5,8 is responsible 
>> for significant noise.  I'll have to be careful with the bias circuit.
>> 
>> Have to get busy for now, but I will report back with results.
>> 
>> Best regards,
>> 
>> Clay
>> 
>>   
> Clay
> 
> One can always use a smaller resistor in series with an RF choke that has no 
> resonances in the region of interest.
> 
> The attached circuit schematic illustrates one method of biasing for which 
> the emitter current of the input transistor can be largely sourced via a 
> resistor rather than from the collector current of the npn output transistor.
> 
> My simulations indicate if that one uses 2N3904's as the input device rather 
> than the 2N5179's shown that there is an enormous peak in the output noise 
> spectrum at around 150-200MHz or so.
> When the 2N5179 is used this noise peak is much smaller and broader.
> 
> Use the same bias divider bypassing techniques that NIST used including the 
> use of electrolytic caps (they used tantalum caps) to reduce the low 
> frequency noise from the power supply. The ceramic bypass caps ensure 
> sufficient isolation between stages.
> Simulating the reverse isolation with realistic component parasitics is 
> always informative/useful.
> 
> Bruce
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Re: [time-nuts] Loran C sounds

[paul swed]

Actually if I had a spare $36M or less I might buy the station on Nantucket.
;-)
Change it from LORAN to Master freq reference...
Though I am thinking several CS standards would be cheaper. Certainly draws
less power

On Thu, Feb 11, 2010 at 10:07 PM, paul swed  wrote:

> Sure they might take a check for $36M or so.
> As they have said without the yanks, its broke. So they will shut down
> before Oct.
> Its all about $$
>
>
> On Thu, Feb 11, 2010 at 9:58 PM, J. Forster  wrote:
>
>> I wonder if there is any point in contacting the Canadians about keeping
>> LORAN operating?
>>
>> -John
>>
>> ===
>>
>>
>> > Unfortunately I had to travel during the shutdown. I had wanted to
>> listen.
>> > But I fired up the austron and 59300 locks (Canadian LORAN)
>> > Like you, I have only ever heard dual rate. Now its a steady patter
>> until
>> > Sept/Oct when even that will go away.
>> >
>> > On Tue, Feb 9, 2010 at 7:47 PM, Rich and Marcia Putz 
>> > wrote:
>> >
>> >> Hi all;
>> >> Just an aside, after hearing dual rated Lorsta Dana for the last 25
>> >> years,
>> >> it is interesting to now hear a single rated chain. Rather than the
>> >> syncopated clatter of Dana, now just a smooth pitter! The east coast
>> >> Canadian chain is much weaker here in northern Indiana than Dana (about
>> >> 125
>> >> miles air) but still quite copyable on my IC-725.
>> >> As far as useability at this point, while Dana was still on I switched
>> >> my
>> >> 2100F over to GRI 5930 and it only took about 1/2 hour to be locked and
>> >> start reading my Austron 1250 frequency numbers. I get my main
>> frequency
>> >> reference from a TAPR T-bolt, but the Austron 2100F was a fun
>> >> experiment. I
>> >> guess the next Loran adventure can be DXing chains.
>> >> Regards; Rich
>> >> ___
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Re: [time-nuts] Loran C sounds

[paul swed]

Sure they might take a check for $36M or so.
As they have said without the yanks, its broke. So they will shut down
before Oct.
Its all about $$

On Thu, Feb 11, 2010 at 9:58 PM, J. Forster  wrote:

> I wonder if there is any point in contacting the Canadians about keeping
> LORAN operating?
>
> -John
>
> ===
>
>
> > Unfortunately I had to travel during the shutdown. I had wanted to
> listen.
> > But I fired up the austron and 59300 locks (Canadian LORAN)
> > Like you, I have only ever heard dual rate. Now its a steady patter until
> > Sept/Oct when even that will go away.
> >
> > On Tue, Feb 9, 2010 at 7:47 PM, Rich and Marcia Putz 
> > wrote:
> >
> >> Hi all;
> >> Just an aside, after hearing dual rated Lorsta Dana for the last 25
> >> years,
> >> it is interesting to now hear a single rated chain. Rather than the
> >> syncopated clatter of Dana, now just a smooth pitter! The east coast
> >> Canadian chain is much weaker here in northern Indiana than Dana (about
> >> 125
> >> miles air) but still quite copyable on my IC-725.
> >> As far as useability at this point, while Dana was still on I switched
> >> my
> >> 2100F over to GRI 5930 and it only took about 1/2 hour to be locked and
> >> start reading my Austron 1250 frequency numbers. I get my main frequency
> >> reference from a TAPR T-bolt, but the Austron 2100F was a fun
> >> experiment. I
> >> guess the next Loran adventure can be DXing chains.
> >> Regards; Rich
> >> ___
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> >
>
>
>
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Re: [time-nuts] Loran C sounds

[J. Forster]

I wonder if there is any point in contacting the Canadians about keeping
LORAN operating?

-John

===


> Unfortunately I had to travel during the shutdown. I had wanted to listen.
> But I fired up the austron and 59300 locks (Canadian LORAN)
> Like you, I have only ever heard dual rate. Now its a steady patter until
> Sept/Oct when even that will go away.
>
> On Tue, Feb 9, 2010 at 7:47 PM, Rich and Marcia Putz 
> wrote:
>
>> Hi all;
>> Just an aside, after hearing dual rated Lorsta Dana for the last 25
>> years,
>> it is interesting to now hear a single rated chain. Rather than the
>> syncopated clatter of Dana, now just a smooth pitter! The east coast
>> Canadian chain is much weaker here in northern Indiana than Dana (about
>> 125
>> miles air) but still quite copyable on my IC-725.
>> As far as useability at this point, while Dana was still on I switched
>> my
>> 2100F over to GRI 5930 and it only took about 1/2 hour to be locked and
>> start reading my Austron 1250 frequency numbers. I get my main frequency
>> reference from a TAPR T-bolt, but the Austron 2100F was a fun
>> experiment. I
>> guess the next Loran adventure can be DXing chains.
>> Regards; Rich
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Re: [time-nuts] WWVB-Style Signal Transmitter in Circuit Cellar

[paul swed]

Great I built up a 60KC generator for testing my wwvb recvrs. It would be
reasonable to add irig B to it. But sounds like ed already is.

On Wed, Feb 10, 2010 at 12:38 AM,  wrote:

> Feb 2010Circuit Cellar (now purchased by Elektor Magazine) has an article
> by Ed Nisley on building a Totally Featureless Clock.  Part 1 describes
> building a WWVB Simulator.  Guess I should build one to take to Dayton and
> leave as a beacon?  N0UU
>
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Re: [time-nuts] Loran C sounds

[paul swed]

Unfortunately I had to travel during the shutdown. I had wanted to listen.
But I fired up the austron and 59300 locks (Canadian LORAN)
Like you, I have only ever heard dual rate. Now its a steady patter until
Sept/Oct when even that will go away.

On Tue, Feb 9, 2010 at 7:47 PM, Rich and Marcia Putz  wrote:

> Hi all;
> Just an aside, after hearing dual rated Lorsta Dana for the last 25 years,
> it is interesting to now hear a single rated chain. Rather than the
> syncopated clatter of Dana, now just a smooth pitter! The east coast
> Canadian chain is much weaker here in northern Indiana than Dana (about 125
> miles air) but still quite copyable on my IC-725.
> As far as useability at this point, while Dana was still on I switched my
> 2100F over to GRI 5930 and it only took about 1/2 hour to be locked and
> start reading my Austron 1250 frequency numbers. I get my main frequency
> reference from a TAPR T-bolt, but the Austron 2100F was a fun experiment. I
> guess the next Loran adventure can be DXing chains.
> Regards; Rich
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Re: [time-nuts] The Smell of Tantalum in the Morning

[paul swed]

My experience is that the plastic ones tend to burn.
The mil grade ones in HP I have never seen fail (As in flame)

On Mon, Feb 8, 2010 at 8:05 PM, Glenn Little WB4UIV <
glennmaill...@bellsouth.net> wrote:

> While in the US Navy, we had to do equipment inspections.
> One quarterly was to examine the capacitors in the power supply of one
> piece of equipment.
> We were to look for leakage (sulfuric acid) from CL65 type wet slug
> tantalum capacitors.
> Shortly after that CL65 type capacitors were disapproved for military use.
> I never saw one that leaked in that equipment, but, have seen a number of
> boards damaged from seal leakage on CL65 capacitors.
> Something to look out for.
> The CL65 capacitors probably have a pure silver case a sulfuric acid as an
> electrolyte.
> The seal is Teflon.
>
> We also had an interesting failure mode for ATC ceramic capacitors.
> This failure mode will only occur in a sealed environment (submarine.
>
> Just an observation.
>
> 73
> Glenn
> WB4UIV
>
>
>
> At 09:46 AM 2/8/2010, you wrote:
>
>> The history of tantalum failures is wide and varied, but
>> there are some common characteristics:
>>
>> 1) The tantalum is in a power supply circuit and receives
>>   a rapid ramp from 0V to operating voltage.
>> 2) The tantalum is spec'd close to its operating voltage,
>>   very close 5V on a 6.3V part, 12.5V on a 15V part...
>> 3) The tantalum is dry slug, and is sealed with epoxy.
>> 4) The instrument has been powered down for an extended
>>   period.
>>
>> HP equipment from the 1980's is pretty immune to the problem
>> because they typically use hermetically sealed mil spec
>> tantalum capacitors.  Tektronix equipment from the 1980's
>> is infested with tantalum problems because they used the
>> cheap epoxy dipped parts.
>>
>> Tantalum failures are pretty rare in equipment that is
>> run continuously.  Tantalum has a self healing feature that
>> corrects any small problems while in operating... Large problems
>> result in detonation.
>>
>> Dipped tantalum capacitors of any age are prone to failure.
>> The tendency can be mitigated largely by never allowing a
>> tantalum capacitor to see voltage above 50% of its rating.
>>
>> And finally, powering a tantalum in reverse, will cause instant
>> and irreparable damage.
>>
>> -Chuck Harris
>>
>>
>>
>> Tom Van Baak wrote:
>>
>>> I powered up a 5071A to watch the end of Loran-C today
>>> and was greeted by the special smell that only a mother
>>> board could love.
>>> Does anyone know the history of tantalum capacitor
>>> failures in ten-year old [HP/Agilent] test equipment?
>>> This is not my first. Last one was more like July 4th.
>>> Thanks,
>>> /tvb
>>>
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[time-nuts] Jupiter GPSDO

[ashley40]

Hello. We built the " 10mhz simple GPSDO from boards supplied by G3RUH. 
(Using the Jupiter GPS receiver)
Questions:
Has anyone built one, 
How well does it work 
How exactly is the RS232 output connected to a XP computer ?
Is the RS323 /software protocol the only way to tell if this thing is actually 
working ?
We are using a trimble ( new) active antenna, mounted with a clear view of the 
sky. 
Does this Jpiter receiver need to be reset before use ? 
How do we tell if we are in a location that is in view of satellites ? 

Reply off list is just fine...  The truth is, we are a bit frustrated with all 
of this gps business, and wonder if its worth any more time . Sorry for the 
long list of grumblings. 
 



 
 
Thank You
Kiss-Electronics
Ms Ashley Hall
183 N 5th Avenue
Cornelius, Oregon
97113
 
 
W7DUZ
 
 
www.kiss-electronics.com
=
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Re: [time-nuts] Tight PLL Tester

[Bruce Griffiths]

WarrenS wrote:


ws said
if what is said does not agree with my experimental results, I'll 
comment.


Bruce said:

0) If one follows that diagram blindly ...

Then that one should not be BUILDING the TESTER from scratch.

1a) The PLL  BW... has to be adjusted to be close to that of the 
oscillator under test.
The PLL Close loop BW is NOWHERE near the Osc freq. I agree that would 
cause severe problems.
The PLL BW has to be high compared to the 1sec LP filter value they 
are using.
What I have shown in the markup, is a 1K to 10 KHz PLL bandwidth which 
makes a good typical value.

IT IS NOT CRITICAL,  just has to be high compared to the low pass filter.


1b) so that the phase detector operates in its linear region.
The Phase detector is ALWAYS operating in its linear region, and NEVER 
off by even a mv at its output.
The high bandwidth 10,000 gain amps between it and the ref osc will 
see to that.



1c) A second order PLL may be a better choice.
A first order works fine at this bandwidth, with NO freq control RCs 
in the loop.

When done right, there is only the natural pole of freq to phase



2 & 3) The mixer LP filter is far from optimum

Which is why I use a C R C R C for mine.
My R's are 49 ohm, Nothing magic about the value, just that with 100 
ohms, they added more Johnson noise.


All such details are important, even the type of resistor and at high 
frequencies the actual part no and manufacturer.
Component parasitics (capacitor inductance, capacitor esr, resistor 
inductance, resistor shunt capacitance, etc)can be very important in RF 
circuits.
Not all resistors (or capacitors) of the same style/size from different 
manufacturers have the same parasitics.



4) One cannot substitute either a DVM or an oversampling ADC
If all that is wanted is a chart recorder output, You can use any DVM 
as shown in the block.
As long as the oversamping ADC is fast compared to the Low pass 1 sec 
filter used in their block
Then the system FREQ noise spectral response as recorded in the PC Log 
file is just about totally determined by the Low pass filter
and NOT the freq response or type of ADC or VtoF converter used or its 
update rate.


I think we're making progress, I didn't see any mention of the 
nonexistent Phase recovering integrator this time.


It should always be present either explicitly or implicitly, to believe 
otherwise is to misunderstand the relationship between average frequency 
and phase differences.


Without an integrator or its equivalent (implicit or explicit) the phase 
noise transfer function will differ from that of the NBS implementation.
One way to approximate a box car integrator is to average the output 
samples in blocks of N samples where N is the number of samples in the 
minimum vale of Tau.
The minimum value of Tau should be around half the RC low pass filter 
cutoff frequency period (you need to read the Stein paper for details).
However the sampling rate has to be sufficiently high (well above the 
Nyquist limit) to allow this without using WSK interpolation to estimate 
the signal value between the samples.

thanks, it's always fun to read your comments
ws

*

Bruce
- Original Message - From: "Bruce Griffiths" 

To: "Discussion of precise time and frequency measurement" 


Sent: Thursday, February 11, 2010 4:02 PM
Subject: Re: [time-nuts] Tight PLL Tester


If one follows that diagram blindly one will encounter a few problems 
with a 10MHz mixer/phase detector input frequency.


1) The PLL is a first order loop and the frequency of the OCXO being 
servoed to the oscillator under test has to be carefully adjusted to 
be close to that of the oscillator under test so that the phase 
detector operates in its linear region. A second order PLL may be a 
better choice.


2) The mixer IF port termination is far from optimum (see later NIST 
papers).
The phase detector sensitivity is much lower than with a better IF 
termination network.
A simple simulation (or test on an actual mixer/phase detector) will 
show this.


3) An off the shelf 750uH inductor will typically exhibit several 
series and parallel resonances in the 100kHz to 20MHz region.
Thus there may still be significant RF at the input of the dc 
amplifier with 80dB gain.
There will be a significant sum frequency (20MHz) component at the 
input to the LC filter.

The dc amplifier following the filter will rectify any RF at its input.
Amplifiers with FET input stages are less sensitive to RF.
An inductor with no resonances below 20MHz is preferred.
100uH inductors with a first SRF greater than 20MHz are available but 
from Germany.
It is usually advisable to use an RC filter between the LC filter 
output and the amplifier input to reduce the RF amplitude seen by the 
dc amplifier.
Another option is to use a cascaded set of passive RC filters instead 
of the LC filter, but this inevitably increases the noise.


4) One cannot substitute either a DVM or an oversampling ADC for the 
V to F converter and 

Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[Bruce Griffiths]

life speed wrote:

Message: 2
Date: Fri, 12 Feb 2010 12:12:29 +1300
From: Bruce Griffiths
The output (collectors of Q5, Q6 emitter of Q4) of the input amplifier
sets the dc voltage at the inputs ( Q1 base, Q7 base respectively) of
the output amplifiers.

The circuit consists of a unity gain input amplifier (Q4, Q5, Q6) that
drives a pair of output amplifiers (Q1, Q2, Q3 and Q7, Q8, Q9
respectively) each with a gain of 2x (6dB).
The input amplifier is essentially a white emitter follower with a
complementary symmetry output stage (shown in transistor electronics
books from the 1960's) where an input CE transistor drives a
complementary pair of CE transistors with feedback from the common
collectors of the 2 output transistors to the input transistor emitter.
In effect its merely a very simple unity gain opamp. Its usually best to
ensure that the CE output stage pair provide the dominant open loop
pole. Using a higher ft (2 to 3x)  input transistor than the output pair
is the usual way of ensuring this.

Well, it is so obvious now that you explained it.  I had forgot about the need 
for one of the stages to set the dominant pole.

Thanks Bruce and Bob for sharing your obsession with frequency controls.  I'll 
simulate this further, and have a prototype PCB built within the next few 
weeks.  I did notice the resistor at the base of Q2,5,8 is responsible for 
significant noise.  I'll have to be careful with the bias circuit.

Have to get busy for now, but I will report back with results.

Best regards,

Clay

   

Clay

One can always use a smaller resistor in series with an RF choke that 
has no resonances in the region of interest.


The attached circuit schematic illustrates one method of biasing for 
which the emitter current of the input transistor can be largely sourced 
via a resistor rather than from the collector current of the npn output 
transistor.


My simulations indicate if that one uses 2N3904's as the input device 
rather than the 2N5179's shown that there is an enormous peak in the 
output noise spectrum at around 150-200MHz or so.

When the 2N5179 is used this noise peak is much smaller and broader.

Use the same bias divider bypassing techniques that NIST used including 
the use of electrolytic caps (they used tantalum caps) to reduce the low 
frequency noise from the power supply. The ceramic bypass caps ensure 
sufficient isolation between stages.
Simulating the reverse isolation with realistic component parasitics is 
always informative/useful.


Bruce
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Re: [time-nuts] Tight PLL Tester

[WarrenS]

ws said
if what is said does not agree with my experimental results, I'll 
comment.


Bruce said:

0) If one follows that diagram blindly ...

Then that one should not be BUILDING the TESTER from scratch.

1a) The PLL  BW... has to be adjusted to be close to that of the 
oscillator under test.
The PLL Close loop BW is NOWHERE near the Osc freq. I agree that would cause 
severe problems.
The PLL BW has to be high compared to the 1sec LP filter value they are 
using.
What I have shown in the markup, is a 1K to 10 KHz PLL bandwidth which makes 
a good typical value.

IT IS NOT CRITICAL,  just has to be high compared to the low pass filter.


1b) so that the phase detector operates in its linear region.
The Phase detector is ALWAYS operating in its linear region, and NEVER off 
by even a mv at its output.
The high bandwidth 10,000 gain amps between it and the ref osc will see to 
that.



1c) A second order PLL may be a better choice.
A first order works fine at this bandwidth, with NO freq control RCs in the 
loop.

When done right, there is only the natural pole of freq to phase



2 & 3) The mixer LP filter is far from optimum

Which is why I use a C R C R C for mine.
My R's are 49 ohm, Nothing magic about the value, just that with 100 ohms, 
they added more Johnson noise.




4) One cannot substitute either a DVM or an oversampling ADC
If all that is wanted is a chart recorder output, You can use any DVM as 
shown in the block.
As long as the oversamping ADC is fast compared to the Low pass 1 sec filter 
used in their block
Then the system FREQ noise spectral response as recorded in the PC Log file 
is just about totally determined by the Low pass filter
and NOT the freq response or type of ADC or VtoF converter used or its 
update rate.


I think we're making progress, I didn't see any mention of the nonexistent 
Phase recovering integrator this time.


thanks, it's always fun to read your comments
ws

*
- Original Message - 
From: "Bruce Griffiths" 
To: "Discussion of precise time and frequency measurement" 


Sent: Thursday, February 11, 2010 4:02 PM
Subject: Re: [time-nuts] Tight PLL Tester


If one follows that diagram blindly one will encounter a few problems with 
a 10MHz mixer/phase detector input frequency.


1) The PLL is a first order loop and the frequency of the OCXO being 
servoed to the oscillator under test has to be carefully adjusted to be 
close to that of the oscillator under test so that the phase detector 
operates in its linear region. A second order PLL may be a better choice.


2) The mixer IF port termination is far from optimum (see later NIST 
papers).
The phase detector sensitivity is much lower than with a better IF 
termination network.
A simple simulation (or test on an actual mixer/phase detector) will show 
this.


3) An off the shelf 750uH inductor will typically exhibit several series 
and parallel resonances in the 100kHz to 20MHz region.
Thus there may still be significant RF at the input of the dc amplifier 
with 80dB gain.
There will be a significant sum frequency (20MHz) component at the input 
to the LC filter.

The dc amplifier following the filter will rectify any RF at its input.
Amplifiers with FET input stages are less sensitive to RF.
An inductor with no resonances below 20MHz is preferred.
100uH inductors with a first SRF greater than 20MHz are available but from 
Germany.
It is usually advisable to use an RC filter between the LC filter output 
and the amplifier input to reduce the RF amplitude seen by the dc 
amplifier.
Another option is to use a cascaded set of passive RC filters instead of 
the LC filter, but this inevitably increases the noise.


4) One cannot substitute either a DVM or an oversampling ADC for the V to 
F converter and counter and produce a set of output samples that will 
necessarily allow one to calculate accurate values for ADEV without 
correcting for the fact that the system phase noise spectral response will 
differ from that when a VFC is used.


If the shape of the phase noise transfer functions differ from that when a 
VFC is used, the computed frequency stability measures obtained will not 
be ADEV, MDEV etc.


Bruce

WarrenS wrote:


Thanks to the persistence and comments of others,
I have marked up an old  NBS diagram to show, anyone that wants to learn, 
how the Tight Phase lock method works to do its 'Magic'.
Although it can be very simple and cheap to build, It does take a certain 
amount of low noise design skill to be able to throw a bunch of parts bin 
things together and make it work as well as it is capable of.
I do believe this information is enough for a well qualified person to 
duplicate or even better my results.

I'm happy to try and answer any specific questions.

also see word discription from:
Page 170 of 'NBS special publication 140' at:
http://digicoll.manoa.hawaii.edu/techreports/PDF/NBS140.pdf81 meg, 
473+ pages  (Takes a while to download)


For another block diagram and short 

Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[life speed]

Message: 2
Date: Fri, 12 Feb 2010 12:12:29 +1300
From: Bruce Griffiths 
The output (collectors of Q5, Q6 emitter of Q4) of the input amplifier 
sets the dc voltage at the inputs ( Q1 base, Q7 base respectively) of 
the output amplifiers.

The circuit consists of a unity gain input amplifier (Q4, Q5, Q6) that 
drives a pair of output amplifiers (Q1, Q2, Q3 and Q7, Q8, Q9 
respectively) each with a gain of 2x (6dB).
The input amplifier is essentially a white emitter follower with a 
complementary symmetry output stage (shown in transistor electronics 
books from the 1960's) where an input CE transistor drives a 
complementary pair of CE transistors with feedback from the common 
collectors of the 2 output transistors to the input transistor emitter. 
In effect its merely a very simple unity gain opamp. Its usually best to 
ensure that the CE output stage pair provide the dominant open loop 
pole. Using a higher ft (2 to 3x)  input transistor than the output pair 
is the usual way of ensuring this.

Well, it is so obvious now that you explained it.  I had forgot about the need 
for one of the stages to set the dominant pole.

Thanks Bruce and Bob for sharing your obsession with frequency controls.  I'll 
simulate this further, and have a prototype PCB built within the next few 
weeks.  I did notice the resistor at the base of Q2,5,8 is responsible for 
significant noise.  I'll have to be careful with the bias circuit.

Have to get busy for now, but I will report back with results.

Best regards,

Clay


  

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Re: [time-nuts] Tight PLL Tester

[Bruce Griffiths]

If one follows that diagram blindly one will encounter a few problems 
with a 10MHz mixer/phase detector input frequency.


1) The PLL is a first order loop and the frequency of the OCXO being 
servoed to the oscillator under test has to be carefully adjusted to be 
close to that of the oscillator under test so that the phase detector 
operates in its linear region. A second order PLL may be a better choice.


2) The mixer IF port termination is far from optimum (see later NIST 
papers).
The phase detector sensitivity is much lower than with a better IF 
termination network.
A simple simulation (or test on an actual mixer/phase detector) will 
show this.


3) An off the shelf 750uH inductor will typically exhibit several series 
and parallel resonances in the 100kHz to 20MHz region.
Thus there may still be significant RF at the input of the dc amplifier 
with 80dB gain.
There will be a significant sum frequency (20MHz) component at the input 
to the LC filter.

The dc amplifier following the filter will rectify any RF at its input.
Amplifiers with FET input stages are less sensitive to RF.
An inductor with no resonances below 20MHz is preferred.
100uH inductors with a first SRF greater than 20MHz are available but 
from Germany.
It is usually advisable to use an RC filter between the LC filter output 
and the amplifier input to reduce the RF amplitude seen by the dc amplifier.
Another option is to use a cascaded set of passive RC filters instead of 
the LC filter, but this inevitably increases the noise.


4) One cannot substitute either a DVM or an oversampling ADC for the V 
to F converter and counter and produce a set of output samples that will 
necessarily allow one to calculate accurate values for ADEV without 
correcting for the fact that the system phase noise spectral response 
will differ from that when a VFC is used.


If the shape of the phase noise transfer functions differ from that when 
a VFC is used, the computed frequency stability measures obtained will 
not be ADEV, MDEV etc.


Bruce

WarrenS wrote:


Thanks to the persistence and comments of others,
I have marked up an old  NBS diagram to show, anyone that wants to 
learn, how the Tight Phase lock method works to do its 'Magic'.
Although it can be very simple and cheap to build, It does take a 
certain amount of low noise design skill to be able to throw a bunch 
of parts bin things together and make it work as well as it is capable 
of.
I do believe this information is enough for a well qualified person to 
duplicate or even better my results.

I'm happy to try and answer any specific questions.

also see word discription from:
Page 170 of 'NBS special publication 140' at:
http://digicoll.manoa.hawaii.edu/techreports/PDF/NBS140.pdf81 meg, 
473+ pages  (Takes a while to download)


For another block diagram and short description also see Figure 1.7 at:
http://tf.nist.gov/phase/Properties/one.htm#oneone

Have Fun
ws

** edited **

Tom

Things will turn out much better to do it the other way around.
When I find out who is going to build/test it,
I'll  make something specific for them that will allow them to be 
able to use there own parts and tool box.


ws

**
- Original Message - From: "Tom Van Baak" 

If there are any Nuts out there interested in helping to make 
available to other Freq-Nuts a SIMPLE tester that I have found to 
be a VERY useful low cost tool,


Warren,

Yes, I think it's a good idea for a couple of people to try to
duplicate your results; either to validate the resolution and
features that you're claiming, or to locate or quantify the
limitations in your implementation. Either way it will be a
learning experience for you, and for the group.

To that end, would you be able this week to write a quick
word document or readme or web page with photo(s) of
your setup, schematic, parts list, specific make/model of
the equipment that you're using, etc. Since you say it is
a simple setup, I suspect a number of us would then be
able to dig in our parts bin and mimic your prototype
as close as possible and then objectively measure how
it works compared to other phase noise measurement
systems.

/tvb









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Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[Bruce Griffiths]

life speed wrote:

Message: 1
Date: Thu, 11 Feb 2010 12:42:27 -0500
From: "Bob Camp"
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
 amplifier(Clay)

Hi

I really should learn how to read the whole message 



Cancel the second request on vibe info.

-

The gotcha with vibration isolation is that it will stop working at some
lower frequency. Aircraft have plenty of vibration running around at low
frequencies.

That all sounds like bad news. Actually it's not. Since the phase noise
isn't going to be all that good below the cutoff of the isolation, the amp
doesn't need to sweat super low phase noise very close in. That can make the
choice of transistors easier.

Bob

Thanks Bob.  I am aware of all the vibe issues, low freq corner, resonance 
peaking, etc.  And yes, I have seen the Wenzel spreadsheet.  Wenzel is a good 
resource for info.  These issues have all been looked into.  The phase noise 
numbers are what is predicted under vibration (10 Hz number might degrade a few 
dB).  The amplifier will need to be better.

1 Hz<  -100 dBc/Hz
10 Hz<  -125 dBc/Hz
100 Hz<  -140 dBc/Hz
1 KHz<  -150 dBc/Hz
10 KHz<  -155 dBc/Hz

Are you aware of any bipolars that are better than others in 1/F noise 
performance?  I noticed Gerhard Hoffman's design used BFG198 and BFG31, 
although those are SOT223 parts, which are somewhat large for my design.  If 
I'm not mistake 'low saturation' correlates to low 1/F noise . . .

I simulated the circuit with two outputs you sent in .GIF format.  It appears 
to be tuned to a somewhat lower frequency than 10 MHz, perhaps 10 KHz to 1 MHz 
where the overall gain is near 0 dB, and the phase shift is near 0.  I am using 
MMBT3904 transistors with Ft near 250 MHz.  Perhaps that is the issue.

Clay



   
The input npn transistors for each stage have a collector current of 
around 2mA the ft of a 2N3904 is relatively low at such currents.
Use a transistor with a higher ft (at 2mA) like a 2N5179 or its SMT 
equivalent for the input transistor.

Increasing the collector current of the input transistor will also help.

There will be a nonzero phase shift at 10MHz due to the finite bandwidth 
of the transistors used.

To a first approximation the phase shift is equivalent to a fixed delay.
This phase shift is relatively unimportant and should have a low tempco.
The gain of the amplifier is more important.
LTSpice predicts an output distortion below -40dBc with a 10MHz input 
and +10dBm output.


Bruce



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Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[Bob Camp]

Hi

If you dig into the data on some of the conventional isolation amp designs, 
their noise floors are down around -150 dbc / Hz in the 1 to 10 Hz range. 
Taking out the transformers and going to a push pull output might bump things 
up into the -140's in the 1 to 10 Hz range.  Essentially you are building the 
"old NIST amp" in the paper:

http://tf.boulder.nist.gov/general/pdf/498.pdf

That's not to say that the old NIST amp is the best way to go, only that the 
close in noise performance should be similar to that amplifier. 

Simple answer = the 2N3904 / 2N3906 should be fine for what you are trying to 
do.  You can find lower noise parts that don't have enough Ft to be useful. You 
can also find RF parts that have great Ft, great broadband noise, and lousy 
flicker noise. 

Bob


On Feb 11, 2010, at 5:57 PM, life speed wrote:

> Message: 1
> Date: Thu, 11 Feb 2010 12:42:27 -0500
> From: "Bob Camp" 
> Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
> amplifier(Clay)
> 
> Hi
> 
> I really should learn how to read the whole message 
> 
>  
> 
> Cancel the second request on vibe info.
> 
> -
> 
> The gotcha with vibration isolation is that it will stop working at some
> lower frequency. Aircraft have plenty of vibration running around at low
> frequencies. 
> 
> That all sounds like bad news. Actually it's not. Since the phase noise
> isn't going to be all that good below the cutoff of the isolation, the amp
> doesn't need to sweat super low phase noise very close in. That can make the
> choice of transistors easier.
> 
> Bob
> 
> Thanks Bob.  I am aware of all the vibe issues, low freq corner, resonance 
> peaking, etc.  And yes, I have seen the Wenzel spreadsheet.  Wenzel is a good 
> resource for info.  These issues have all been looked into.  The phase noise 
> numbers are what is predicted under vibration (10 Hz number might degrade a 
> few dB).  The amplifier will need to be better.
> 
> 1 Hz < -100 dBc/Hz
> 10 Hz < -125 dBc/Hz
> 100 Hz < -140 dBc/Hz
> 1 KHz < -150 dBc/Hz
> 10 KHz < -155 dBc/Hz
> 
> Are you aware of any bipolars that are better than others in 1/F noise 
> performance?  I noticed Gerhard Hoffman's design used BFG198 and BFG31, 
> although those are SOT223 parts, which are somewhat large for my design.  If 
> I'm not mistake 'low saturation' correlates to low 1/F noise . . .
> 
> I simulated the circuit with two outputs you sent in .GIF format.  It appears 
> to be tuned to a somewhat lower frequency than 10 MHz, perhaps 10 KHz to 1 
> MHz where the overall gain is near 0 dB, and the phase shift is near 0.  I am 
> using MMBT3904 transistors with Ft near 250 MHz.  Perhaps that is the issue.
> 
> Clay
> 
> 
> 
> 
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> 


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Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier (Clay)

[Bruce Griffiths]

life speed wrote:

Message: 2
Date: Thu, 11 Feb 2010 07:54:40 -0500
From: Bob Camp

Hi

Implementing that circuit without using a hybrid would be a bit of a challenge.

Bob

Message: 6
Date: Fri, 12 Feb 2010 04:09:08 +1300
From: Bruce Griffiths

Yes implementing an exact copy without using a hybrid would be difficult.
However for 10MHz use, its probably not too difficult since that
isolation amplifier is intended for a 100MHz signal and the requirement
is for 10MHz operation.

If the transistor ft's are reduced by a factor of 10 or so it shouldn't
be too much of a problem.
At 10MHz 2N3906 and 2N3904 transistors should suffice.

Bruce

Hi Bruce,

Thanks for the tips.  I've been trying to follow the circuits you posted.  The 
first one, in .PNG format, looks like a common-base complementary (push-pull) 
stage followed by a common-emitter complementary stage to provide the low 
impedance output.

The second circut in .GIF fromat I am having a bit more trouble understanding.  
I see that V6, 7 are at the outputs and just used for to simulate isolation.  
V1 is the input?

V1 is indeed the input.

  Are Q5,6 used to set the bias point of Q4?  Are V2,3,4 just there to bias the 
transistors for simulation purposes, and this would be accomplished another way 
in a real implementation?
   


The output (collectors of Q5, Q6 emitter of Q4) of the input amplifier 
sets the dc voltage at the inputs ( Q1 base, Q7 base respectively) of 
the output amplifiers.


The circuit consists of a unity gain input amplifier (Q4, Q5, Q6) that 
drives a pair of output amplifiers (Q1, Q2, Q3 and Q7, Q8, Q9 
respectively) each with a gain of 2x (6dB).
The input amplifier is essentially a white emitter follower with a 
complementary symmetry output stage (shown in transistor electronics 
books from the 1960's) where an input CE transistor drives a 
complementary pair of CE transistors with feedback from the common 
collectors of the 2 output transistors to the input transistor emitter. 
In effect its merely a very simple unity gain opamp. Its usually best to 
ensure that the CE output stage pair provide the dominant open loop 
pole. Using a higher ft (2 to 3x)  input transistor than the output pair 
is the usual way of ensuring this.




The output stages can be viewed as simple 3 transistor current feedback 
opamps with a nominal gain of about 2x (6dB).
The output stage gain being adjusted in this case for an overall gain of 
0dB when driving a 50 ohm load.
The 47 ohm resistors in series with the outputs match the output 
impedance to that of a 50 ohm cable.
With a closed loop gain of 2 ensuring that the ft of the input 
transistor is greater (>2x) than that of the output stage transistors is 
less critical.


Both output transistors contribute to the RF output signal.
The npn output transistor are also used to set the operating current of 
the output stage.
The resistor in series with the npn output transistor emitter is 
bypassed for RF so that the full gain of this transistor is available at RF.
Using a complementary symmetry output stage allows the dc collector 
current of the output stage to be reduced to about half that required if 
the npn output stage transistor were merely acting as a fixed current 
source.


Yes the 1.7V dc sources are only included for the simulation.
I just wanted to illustrate the principles without getting into too much 
detail in that post.
In practice one could either use a LED or a resistive voltage divider 
buffered by a pnp emitter follower (either method provides a degree of 
temperature compensation for Vbe tempco of the npn output transistors) .
Either one uses independent biasing for each npn CE device, or elaborate 
RC filtering (at least an independent 2 stage filter for each transistor 
) to avoid degrading the RF reverse isolation via the RF impedance of 
the common 1.7V bias circuit.


As noted in a later post, using unity gain output amplifiers with a 2x 
gain input stage allows the total dc current to be reduced below that of 
when the input stage has unity gain and the output stages have voltage 
gain of 2X.

Please explain the comment regarding the hybrid.  Are you and Bob referring to 
a 90 degree hybrid coupler, or other quadrature method like a transmission line 
transformer?  What would be the purpose of such a device?

Would it be too much to ask for a description of these circuits?  I suppose we 
all have our areas of expertise, and transistor isolation amps are somewhat new 
to me.

Thanks again for all the help.

Clay

PS - yes, the OCXO is vibe isolated.  And you are certainly correct about long 
runs of single-ended coax being susceptible to noise.  The system designer has 
accepted this and allowed for some degradation.  But I will look into the 
practicality of implementing a differential line for the long run of 10 MHz 
cable.  However, I will still need to implement traditional coaxial isolated 10 
MHz outputs.

   

Bruce


___

Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[life speed]

Message: 1
Date: Thu, 11 Feb 2010 12:42:27 -0500
From: "Bob Camp" 
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
    amplifier(Clay)

Hi

I really should learn how to read the whole message 

 

Cancel the second request on vibe info.

-

The gotcha with vibration isolation is that it will stop working at some
lower frequency. Aircraft have plenty of vibration running around at low
frequencies. 

That all sounds like bad news. Actually it's not. Since the phase noise
isn't going to be all that good below the cutoff of the isolation, the amp
doesn't need to sweat super low phase noise very close in. That can make the
choice of transistors easier.

Bob

Thanks Bob.  I am aware of all the vibe issues, low freq corner, resonance 
peaking, etc.  And yes, I have seen the Wenzel spreadsheet.  Wenzel is a good 
resource for info.  These issues have all been looked into.  The phase noise 
numbers are what is predicted under vibration (10 Hz number might degrade a few 
dB).  The amplifier will need to be better.

1 Hz < -100 dBc/Hz
10 Hz < -125 dBc/Hz
100 Hz < -140 dBc/Hz
1 KHz < -150 dBc/Hz
10 KHz < -155 dBc/Hz

Are you aware of any bipolars that are better than others in 1/F noise 
performance?  I noticed Gerhard Hoffman's design used BFG198 and BFG31, 
although those are SOT223 parts, which are somewhat large for my design.  If 
I'm not mistake 'low saturation' correlates to low 1/F noise . . .

I simulated the circuit with two outputs you sent in .GIF format.  It appears 
to be tuned to a somewhat lower frequency than 10 MHz, perhaps 10 KHz to 1 MHz 
where the overall gain is near 0 dB, and the phase shift is near 0.  I am using 
MMBT3904 transistors with Ft near 250 MHz.  Perhaps that is the issue.

Clay


  

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[time-nuts] IL waveforms

[Kit Scally]

Joop,

Indeed !  
During lab evaluation, a sine wave locking source required much greater
power (15-20dB IIRC) to achieve lock.  Further, lock limits were almost
impossible to discern with any degree of confidence.  
With a square wave, there were distinctive step jumps when approaching
lock & going out of lock.

Kit
VK2LL

Kit VK2LL wrote:
>Theory suggests that lock sensitivity could be improved in this
instance
>if the 10MHz duty cycle was changed from 1:1 to 6 or 7:1, the idea
being
>that the narrower pulse should be <0.5 period of the frequency of the
>oscillator to be locked.  

Joop wrote:
Thinking this way of injection locking, it makes sense to increase
harmonic content of the injected signal. In essence it will then not
lock
onto the fundamental frequency, but on some harmonic part of it. In my
SPICE investigation I injected a clean sine signal. Perhaps I should try
with some clipped or digital signal.
That might also explain why James G1PVZ was so successful. The bottom
2N918 might cause a lot of harmonics if driven into the collector
cut-off.

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[time-nuts] IL graph

[Kit Scally]

Didier, 

This wasn't the  version I had in mind, but it contains
essentially the same information.  
Thanks for the link.

Kit
***
This link will show you the effect of duty cycle on harmonic content:

http://www.ko4bb.com/Test_Equipment/Pulse_Modulation/

Didier KO4BB



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Re: [time-nuts] [FS] HP 34420A NanoVolt Micro-Ohm cable

[Samuel DEMEULEMEESTER]

I take them here too :D

-Message d'origine-
De : time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com] De la
part de Peter Loron
Envoyé : jeudi 11 février 2010 21:48
À : Discussion of precise time and frequency measurement
Objet : [time-nuts] [FS] HP 34420A NanoVolt Micro-Ohm cable

I have three new HP cables for the 34420A NanoVolt, Micro-Ohm Meter. These
appear to be for connection to a calibration system, but I'm not sure. The
part number on the package is 97173351. These appear to be custom cables for
connecting the meter to some apparatus.

Asking $10 + shipping.

-Pete
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[time-nuts] [FS] HP 34420A NanoVolt Micro-Ohm cable

[Peter Loron]

I have three new HP cables for the 34420A NanoVolt, Micro-Ohm Meter. These 
appear to be for connection to a calibration system, but I'm not sure. The part 
number on the package is 97173351. These appear to be custom cables for 
connecting the meter to some apparatus.

Asking $10 + shipping.

-Pete
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Re: [time-nuts] Vibrations

[Don Latham]

There is a heat conductive rubber gasket material for automotive use...It's 
the higher frequency mechanical thumps you need to get rid of.

Don

- Original Message - 
From: "Stan, W1LE" 
To: "Discussion of precise time and frequency measurement" 


Sent: Thursday, February 11, 2010 7:30 AM
Subject: Re: [time-nuts] Vibrations



Hello Raj,

I integrated a Symmetricom model X72 Rb  into a Down East Microwave (DEMI) 
chassis
and it does experience microphonics. This was for a friends 10 GHz 
transceiver used for portable work.

So getting rid of the microphonics is important.
Somehow I need to mechanically isolate the Rb from the chassis and still 
have a heat sink.


Stan, W1LE


Raj wrote:
While comparing a Rb FE5680 frequency reference with another RB or GPSDO 
I find that tapping a Rb unit causes a sudden shift in the scope display 
meaning the frequency has slightly shifted momentarily and locks back 
steadily with a phase shift. This does not happen in another Rb FE5680.


Would something be loose I wonder ? Any experiences?

Cheers





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Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[Bob Camp]

Hi

There's a quick and dirty phase noise to vibration spread sheet here:

http://www.wenzel.com/documents/spread.htm

You may want to plug in some numbers and see what you get. A lot is going to
depend on your oscillator's sensitivity, vibration isolation setup, and the
operational (as opposed to survival) vibration profile. 

The added noise may take you into an area where a high speed op-amp based
design is quite adequate. Some topologies might also help with generating a
balanced output feed.

One downside to op amps would be power. Most of the circuits that have been
tossed around are already a bit outside the original budget you came up
with. That may or may not be significant in your case.

Bob

-Original Message-
From: time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com] On
Behalf Of life speed
Sent: Thursday, February 11, 2010 1:05 PM
To: time-nuts@febo.com
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)


Message: 6
Date: Thu, 11 Feb 2010 12:38:58 -0500
From: "Bob Camp" 
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
    amplifier(Clay)

Hi

I'll grab the one on the hybrid.

In this case hybrid is referring to a construction technique.

The circuit shown was originally fabricated in a TO-8 package with chip and
wire construction. It was certainly made using thin film or thick film
technology on a substrate. Based on the number of components and size of the
part, I'd bet that the resistors were printed on the substrate. 

When you are using that kind of construction approach there are some good
things that happen and some bad things. The circuit topology is modified to
work with the construction technique. In this case the Ft's of the
transistors are quite high. Taming them on a substrate (alumina or glass) is
a very different thing than doing it on a PC board. 

---

Is your OCXO vibration isolated?

--

Bob

OK, you're talking about chip-and-wire, or hybrid construction.  Really, at
10 MHz, that seems unnecessary.  Of course, PCB construction will introduce
some parasitics.  Transistors with lower Ft could be used.  Additionally,
there are bandwidth-limiting techniques like adding feedback capacitance.  I
suppose this would come at the expense of high-frequency isolation.  But for
my application isolation is important in the 10's of MHz, not much higher. 
Having built a couple transistor circuits over the years, I am aware of what
can happen when a transistor in a low frequency circuit has an Ft of
multiple GHz.  Usually oscillations.  I imagine a high Ft enables the
isolation to extend to very high frequencies.

Yes the OCXO is vibration isolated or the system wouldn't produce good phase
noise in operation.

Clay


  

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Re: [time-nuts] Advice on 10 MHz isolation/distribution (Clay)

[life speed]

Message: 6
Date: Thu, 11 Feb 2010 12:38:58 -0500
From: "Bob Camp" 
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
    amplifier(Clay)

Hi

I'll grab the one on the hybrid.

In this case hybrid is referring to a construction technique.

The circuit shown was originally fabricated in a TO-8 package with chip and
wire construction. It was certainly made using thin film or thick film
technology on a substrate. Based on the number of components and size of the
part, I'd bet that the resistors were printed on the substrate. 

When you are using that kind of construction approach there are some good
things that happen and some bad things. The circuit topology is modified to
work with the construction technique. In this case the Ft's of the
transistors are quite high. Taming them on a substrate (alumina or glass) is
a very different thing than doing it on a PC board. 

---

Is your OCXO vibration isolated?

--

Bob

OK, you're talking about chip-and-wire, or hybrid construction.  Really, at 10 
MHz, that seems unnecessary.  Of course, PCB construction will introduce some 
parasitics.  Transistors with lower Ft could be used.  Additionally, there are 
bandwidth-limiting techniques like adding feedback capacitance.  I suppose this 
would come at the expense of high-frequency isolation.  But for my application 
isolation is important in the 10's of MHz, not much higher.  Having built a 
couple transistor circuits over the years, I am aware of what can happen when a 
transistor in a low frequency circuit has an Ft of multiple GHz.  Usually 
oscillations.  I imagine a high Ft enables the isolation to extend to very high 
frequencies.

Yes the OCXO is vibration isolated or the system wouldn't produce good phase 
noise in operation.

Clay


  

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Re: [time-nuts] Injection locking

[Brooke Clarke]

Hi:

A continuation of the idea of using something other than a square wave 
is what Don Lancaster promoted that were called "Magic Sine Waves".  The 
idea is to come up with a waveform that maximizes the desired output 
sine frequency while greatly suppressing the low order harmonics.  The 
result is that a very simple low pass filter will yield a low distortion 
sine wave. This was aimed at DC to AC inverters, but with high speed 
logic could be used to generate an RF sine wave.


Have Fun,

Brooke Clarke
http://www.PRC68.com


Rex wrote:

Kit Scally wrote:

...

On a related topic, I found some while ago - and promptly lost - a
graph/chart showing harmonic level variations with varying duty-cycle of
an input waveform.  This was to some degree a graphical representation
of the Wenzel document referenced by Bruce recently.  Has anyone got a
link to this document please ?


Kit
VK2LL


Is this the Wenzel document you are referring to?
http://www.wenzel.com/pdffiles1/pdfs/choose.pdf  ("Choosing a 
Frequency Multiplier's Waveform")


Figure 2 in that doc is a graph of Harmonic Amplitudes vs. pulse 
width. Is that what you were looking for? There are a couple of bar 
charts too, for specific duty cycles.





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Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier(Clay)

[Bob Camp]

Hi

I really should learn how to read the whole message 

 

Cancel the second request on vibe info.

-

The gotcha with vibration isolation is that it will stop working at some
lower frequency. Aircraft have plenty of vibration running around at low
frequencies. 

That all sounds like bad news. Actually it's not. Since the phase noise
isn't going to be all that good below the cutoff of the isolation, the amp
doesn't need to sweat super low phase noise very close in. That can make the
choice of transistors easier.

Bob

-Original Message-
From: time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com] On
Behalf Of life speed
Sent: Thursday, February 11, 2010 12:27 PM
To: time-nuts@febo.com
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
amplifier(Clay)

Message: 2
Date: Thu, 11 Feb 2010 07:54:40 -0500
From: Bob Camp 

Hi

Implementing that circuit without using a hybrid would be a bit of a
challenge. 

Bob

Message: 6
Date: Fri, 12 Feb 2010 04:09:08 +1300
From: Bruce Griffiths 

Yes implementing an exact copy without using a hybrid would be difficult.
However for 10MHz use, its probably not too difficult since that 
isolation amplifier is intended for a 100MHz signal and the requirement 
is for 10MHz operation.

If the transistor ft's are reduced by a factor of 10 or so it shouldn't 
be too much of a problem.
At 10MHz 2N3906 and 2N3904 transistors should suffice.

Bruce

Hi Bruce,

Thanks for the tips.  I've been trying to follow the circuits you posted. 
The first one, in .PNG format, looks like a common-base complementary
(push-pull) stage followed by a common-emitter complementary stage to
provide the low impedance output.

The second circut in .GIF fromat I am having a bit more trouble
understanding.  I see that V6, 7 are at the outputs and just used for to
simulate isolation.  V1 is the input? Are Q5,6 used to set the bias point of
Q4?  Are V2,3,4 just there to bias the transistors for simulation purposes,
and this would be accomplished another way in a real implementation?

Please explain the comment regarding the hybrid.  Are you and Bob referring
to a 90 degree hybrid coupler, or other quadrature method like a
transmission line transformer?  What would be the purpose of such a device?

Would it be too much to ask for a description of these circuits?  I suppose
we all have our areas of expertise, and transistor isolation amps are
somewhat new to me.

Thanks again for all the help.

Clay

PS - yes, the OCXO is vibe isolated.  And you are certainly correct about
long runs of single-ended coax being susceptible to noise.  The system
designer has accepted this and allowed for some degradation.  But I will
look into the practicality of implementing a differential line for the long
run of 10 MHz cable.  However, I will still need to implement traditional
coaxial isolated 10 MHz outputs.


  

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Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier(Clay)

[Bob Camp]

Hi

I'll grab the one on the hybrid.

In this case hybrid is referring to a construction technique.

The circuit shown was originally fabricated in a TO-8 package with chip and
wire construction. It was certainly made using thin film or thick film
technology on a substrate. Based on the number of components and size of the
part, I'd bet that the resistors were printed on the substrate. 

When you are using that kind of construction approach there are some good
things that happen and some bad things. The circuit topology is modified to
work with the construction technique. In this case the Ft's of the
transistors are quite high. Taming them on a substrate (alumina or glass) is
a very different thing than doing it on a PC board. 

---

Is your OCXO vibration isolated?

--

Bob

-Original Message-
From: time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com] On
Behalf Of life speed
Sent: Thursday, February 11, 2010 12:27 PM
To: time-nuts@febo.com
Subject: Re: [time-nuts] Advice on 10 MHz isolation/distribution
amplifier(Clay)

Message: 2
Date: Thu, 11 Feb 2010 07:54:40 -0500
From: Bob Camp 

Hi

Implementing that circuit without using a hybrid would be a bit of a
challenge. 

Bob

Message: 6
Date: Fri, 12 Feb 2010 04:09:08 +1300
From: Bruce Griffiths 

Yes implementing an exact copy without using a hybrid would be difficult.
However for 10MHz use, its probably not too difficult since that 
isolation amplifier is intended for a 100MHz signal and the requirement 
is for 10MHz operation.

If the transistor ft's are reduced by a factor of 10 or so it shouldn't 
be too much of a problem.
At 10MHz 2N3906 and 2N3904 transistors should suffice.

Bruce

Hi Bruce,

Thanks for the tips.  I've been trying to follow the circuits you posted. 
The first one, in .PNG format, looks like a common-base complementary
(push-pull) stage followed by a common-emitter complementary stage to
provide the low impedance output.

The second circut in .GIF fromat I am having a bit more trouble
understanding.  I see that V6, 7 are at the outputs and just used for to
simulate isolation.  V1 is the input? Are Q5,6 used to set the bias point of
Q4?  Are V2,3,4 just there to bias the transistors for simulation purposes,
and this would be accomplished another way in a real implementation?

Please explain the comment regarding the hybrid.  Are you and Bob referring
to a 90 degree hybrid coupler, or other quadrature method like a
transmission line transformer?  What would be the purpose of such a device?

Would it be too much to ask for a description of these circuits?  I suppose
we all have our areas of expertise, and transistor isolation amps are
somewhat new to me.

Thanks again for all the help.

Clay

PS - yes, the OCXO is vibe isolated.  And you are certainly correct about
long runs of single-ended coax being susceptible to noise.  The system
designer has accepted this and allowed for some degradation.  But I will
look into the practicality of implementing a differential line for the long
run of 10 MHz cable.  However, I will still need to implement traditional
coaxial isolated 10 MHz outputs.


  

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Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier (Clay)

[life speed]

Message: 2
Date: Thu, 11 Feb 2010 07:54:40 -0500
From: Bob Camp 

Hi

Implementing that circuit without using a hybrid would be a bit of a challenge. 

Bob

Message: 6
Date: Fri, 12 Feb 2010 04:09:08 +1300
From: Bruce Griffiths 

Yes implementing an exact copy without using a hybrid would be difficult.
However for 10MHz use, its probably not too difficult since that 
isolation amplifier is intended for a 100MHz signal and the requirement 
is for 10MHz operation.

If the transistor ft's are reduced by a factor of 10 or so it shouldn't 
be too much of a problem.
At 10MHz 2N3906 and 2N3904 transistors should suffice.

Bruce

Hi Bruce,

Thanks for the tips.  I've been trying to follow the circuits you posted.  The 
first one, in .PNG format, looks like a common-base complementary (push-pull) 
stage followed by a common-emitter complementary stage to provide the low 
impedance output.

The second circut in .GIF fromat I am having a bit more trouble understanding.  
I see that V6, 7 are at the outputs and just used for to simulate isolation.  
V1 is the input? Are Q5,6 used to set the bias point of Q4?  Are V2,3,4 just 
there to bias the transistors for simulation purposes, and this would be 
accomplished another way in a real implementation?

Please explain the comment regarding the hybrid.  Are you and Bob referring to 
a 90 degree hybrid coupler, or other quadrature method like a transmission line 
transformer?  What would be the purpose of such a device?

Would it be too much to ask for a description of these circuits?  I suppose we 
all have our areas of expertise, and transistor isolation amps are somewhat new 
to me.

Thanks again for all the help.

Clay

PS - yes, the OCXO is vibe isolated.  And you are certainly correct about long 
runs of single-ended coax being susceptible to noise.  The system designer has 
accepted this and allowed for some degradation.  But I will look into the 
practicality of implementing a differential line for the long run of 10 MHz 
cable.  However, I will still need to implement traditional coaxial isolated 10 
MHz outputs.


  

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Re: [time-nuts] Injection locking

[Bob Camp]

Hi

If you want to start with sine waves rather than square waves, there is a
lot of data on duty cycle and harmonics. Often you find it with stuff in the
title like "clipped cosine wave". 

Just about any RF design text book from the 1940's or 1950's probably will
have some nice charts in it. Fiddling bias on class C triode amplifiers used
to be a major part of the design. 

No, I will not under any circumstances reveal which course I learned that in
and how long ago it was

Of course a simple Excel spread sheet can come up with all of this pretty
fast.  

Bob

-Original Message-
From: time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com] On
Behalf Of Didier Juges
Sent: Thursday, February 11, 2010 10:42 AM
To: Time-Nuts
Subject: Re: [time-nuts] Injection locking

This link will show you the effect of duty cycle on harmonic content:

http://www.ko4bb.com/Test_Equipment/Pulse_Modulation/

Didier KO4BB

 Sent from my BlackBerry Wireless thingy while I do
other things... 

-Original Message-
From: Rex 
Date: Thu, 11 Feb 2010 02:09:48 
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Injection locking

Kit Scally wrote:
> ...
>
> On a related topic, I found some while ago - and promptly lost - a
> graph/chart showing harmonic level variations with varying duty-cycle of
> an input waveform.  This was to some degree a graphical representation
> of the Wenzel document referenced by Bruce recently.  Has anyone got a
> link to this document please ?
>
>
> Kit
> VK2LL 
>
>   

Is this the Wenzel document you are referring to?
http://www.wenzel.com/pdffiles1/pdfs/choose.pdf  ("Choosing a Frequency 
Multiplier's Waveform")

Figure 2 in that doc is a graph of Harmonic Amplitudes vs. pulse width. 
Is that what you were looking for? There are a couple of bar charts too, 
for specific duty cycles.




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Re: [time-nuts] Vibrations

[Bob Camp]

Hi

The case of a rubidium acts as a magnetic shield. I would not recommend
operating one without the case or any of the internal shields and covers. 

The rubidium physics package operates on the internal crystal oscillator
through a fairly narrow loop bandwidth. If the issue looks like microphonics
I would start looking at the OCXO and buffering first. 

The physics package needs to heat up in order to do it's work. If you try to
cool it, the heaters will just draw more power to keep it hot. 

With any standard, keeping the external environment constant is more
important than keeping it at one temperature or another. The first thing to
do is to eliminate drafts. The stuff will run a few degrees warmer, but it
will be more stable.

Bob

-Original Message-
From: time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com] On
Behalf Of Raj
Sent: Thursday, February 11, 2010 9:47 AM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Vibrations

Hi Stan,

I will explore this issue tomorrow, 20:00 here now. I was just going
through a boyish thrill of fiddling with a Sony ICF-S10Mk2 and the amazing
numbers of AM stations it picks up.. maybe it can be used at 60Khz.. for 10$
it worth it !!

How about keeping he Rb cover open ? I measured the lamp housing at
110C and the rest at 60-70C. I dont intend to keep my units on all the time
but I dont feel good about anything running hot! Maybe a small slow fan
might help. I am fascinated by the blinking light inside at 1Hz and thats
not available on the DB connector. I did add a reverse polarity protection
by soldering a 2A diode between the center pin of the regulator and the +15
pin. I wish I can find a schematic and what else I can do with this gizmo!

73 Raj vu2zap

At 11-02-10, you wrote:
>Hello Raj,
>
>I integrated a Symmetricom model X72 Rb  into a Down East Microwave (DEMI)
chassis
>and it does experience microphonics. This was for a friends 10 GHz
transceiver used for portable work.
>So getting rid of the microphonics is important.
>Somehow I need to mechanically isolate the Rb from the chassis and still
have a heat sink.
>
>Stan, W1LE
>
>
>Raj wrote:
>>While comparing a Rb FE5680 frequency reference with another RB or GPSDO I
find that tapping a Rb unit causes a sudden shift in the scope display
meaning the frequency has slightly shifted momentarily and locks back
steadily with a phase shift. This does not happen in another Rb FE5680.
>>
>>Would something be loose I wonder ? Any experiences?
>>
>>Cheers
>>
>>  
>
>
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>time-nuts mailing list -- time-nuts@febo.com
>To unsubscribe, go to
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>and follow the instructions there.

-- 
Raj, VU2ZAP
Bangalore, India. 


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Re: [time-nuts] Injection locking

[Didier Juges]

This link will show you the effect of duty cycle on harmonic content:

http://www.ko4bb.com/Test_Equipment/Pulse_Modulation/

Didier KO4BB

 Sent from my BlackBerry Wireless thingy while I do 
other things... 

-Original Message-
From: Rex 
Date: Thu, 11 Feb 2010 02:09:48 
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Injection locking

Kit Scally wrote:
> ...
>
> On a related topic, I found some while ago - and promptly lost - a
> graph/chart showing harmonic level variations with varying duty-cycle of
> an input waveform.  This was to some degree a graphical representation
> of the Wenzel document referenced by Bruce recently.  Has anyone got a
> link to this document please ?
>
>
> Kit
> VK2LL 
>
>   

Is this the Wenzel document you are referring to?
http://www.wenzel.com/pdffiles1/pdfs/choose.pdf  ("Choosing a Frequency 
Multiplier's Waveform")

Figure 2 in that doc is a graph of Harmonic Amplitudes vs. pulse width. 
Is that what you were looking for? There are a couple of bar charts too, 
for specific duty cycles.




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Re: [time-nuts] TBolt self survey

[Raj]

Thanks for all the info Arnold. I will study all these links ASAP.

I had a Garmin 12A (many years ago) that gave the actual height close to the 
survey value of 928M ASL. I wonder if those calculated the real values..


>Raj,
>
>don't trust the height asl from (older) maps. There are so many different 
>local 
>sea level definitions around the world, which are not yet WGS84/ EGM96 
>conform (WGS 1972 would be good as well)! 
>I had similar problems comparing swiss and and german map values 
>with the Shuttle Radar Topography Mission (SRTM 2000) values.
>
>On http://www.colorado.edu/geography/gcraft/notes/datum/edlist.html 
>I found interesting informations about the differences (tables).
>Eg. India used the Everest ellipsoid (India 1956, for Inda and Nepal) 
>where the differences to WGS84 are (d = delta): 
>dX= 295, dY= 736, dZ= 257 meters, what might explain already some 
>discrepancies. Attention to the horizontal displacements as well!
>
>So nothing to worry, as already correctly explained by several other 
>fellow Time Nuts, use the elevation found with the auto position function
>in LH, this is the right GPS value for this purpose ( perhaps moving around 
>up to 1 m or similar), which is of no concern I think. Mark Sims and 
>John Miles did really a good job! 
>(The self survey function in Thunderbolt Monitor does give a similar 
>result with somewhat less resolution/ precision.)
>
>Searching around I found another page with good informations for 
>beginners: 'Datum of a GPS Receiver'
>http://homepages.slingshot.co.nz/~geoff36/datum.htm
>
>The therein mentioned calculator 
>'Geoid Height Calculator NGA EGM96', url:
>http://earth-info.nga.mil/GandG/wgs84/gravitymod/egm96/intpt.html
>does give the same results...
>
>enjoy the time
>
>Arnold, DK2WT
>
>On Thu, 11 Feb 2010 06:26:42 +0530, Raj wrote:
>
>>My GPS reading is about 850 +- 1
>>The "official" height ASL = 920
>
>>This calculator gives me ~936
>
>>So what should I use ?
>
>>Thanks
>>Raj
>
>>At 10-02-10, you wrote:
>>>Try to find your correct height here:
>>>http://sps.unavco.org/geoid/
>>>
>>>If you put in your correct coordinates without any input for the elevation, 
>>>you will get theorthometric height, which is the GPS ellipsoidal height 
>>>minus geoid height. That is the amount you need to subtract from 
>>>your Thunderbolt GPS height in order to get your local geodetic height 
>>>(WGS84). 
>>>
>>>Filling in the Thunderbolt GPS height you will get the 'real' height 
>>>for your antenna...
>>>
>>>Some explanations:
>>>http://www.unavco.org/edu_outreach/tutorial/geoidcorr.html
>>>
>>>On my location I have always to reduce the GPS height by 46.623 m.
>>>Good luck with TB and the Lady Heather,
>>>
>>>Arnold, DK2WT
>>>
>>>
>>>
>>>On Wed, 10 Feb 2010 14:20:36 +0100, EB4APL wrote:
>>>
Raj,
>>>
Ask them about ellipsoidal and orthometric heights, or the local 
difference between the geoid and the WGS84 ellipsoid.  Also your long 
and lat may change depending of the Geodetic Reference System chosen.  
By default GPS receivers use WGS84 and ellipsoidal height.
>>>
73 de Ignacio, EB4APL
>>>
--
Raj wrote:
> Thanks Stan,
>
> That worked, I did not see the  control menu properly!
>
> I set the co-ords to 0 0 0 and started survey. The Tbolt found the Lat 
> long withing a few minutes and only the altitude is about 70 M below the 
> official figure for my place. I live a a few hundred meters from 
> Bangalore center and I am at the same height.. must check with the survey 
> dept. down the road!!
>
> 73
> Raj vu2zap
>
>
> At 10-02-10, you wrote:
>   
>> Hello Raj,
>>
>> using the trimble T'boltMON software:
>>
>> Start with a "factory reset" in the control menu
>>
>> follow previous guidance of enabling the "save position" capability,
>>
>> Then to restart a survey
>>
>> go to the control menu (on left),
>> select "restart self survey"
>>
>>
>> In lady Heather V3.0 beta of 21 Jan '10 ( available from the KE5FX 
>> website):
>> keyboard enter a "s", for survey
>> then "s" for a standard survey,
>> use the default of 2000 samples,  for a start
>> Then "enter"
>>
>> Time for me to go to bed.
>>
>> Please advise of progress.
>>
>> Stan,  W1LE
>>
>>
>> Raj wrote:
>> 
>>> Thanks Stan,
>>>
>>> I did that but I can't where to "start the survey" in TBolt. In lady 
>>> heather there
>>> is a key for it.
>>>
>>> At 09-02-10, you wrote:
>>>  
>>>   
 Hello Raj,

 that location is probably where the coordinates were last saved. or 
 there abouts
 Or those are the default factory settings from memory,
 and new coordinates were never saved.

 redo the survey and save

Re: [time-nuts] Tight PLL Tester

[Bruce Griffiths]

WarrenS wrote:

Bruce

Thanks for your response, as always you've give me plenty to think about.

Bruce said:  It is essential to understand exactly how this system 
works in theory.

Turns out to be too true.
After re-reading your last post several times,
I now finally understand what you are saying and why you are saying it.
It is because YOU do not yet understand how this method works.
I find that so unbelievable, that I had not considered that possibility.

Starting at the following line and pretty much everything after that, 
although accurate statements,

THEY DO NOT APPLY to this method.



To recover the phase fluctuations the EFC voltage has to be integrated.

...

ws


Warren

You seem to be the only one who doesn't understand the theory.
I understand exactly how the method as implemented by NBS is intended to 
work.

Your ad hoc assumptions about the details of the method are false.

You admit to not knowing how to calculate how your implementation 
responds to different phase noise spectra and yet you confidently 
proclaim there will be no problems in interpreting the results?


Bruce

***
- Original Message - From: "Bruce Griffiths" 

To: "Discussion of precise time and frequency measurement" 


Sent: Wednesday, February 10, 2010 2:32 PM
Subject: Re: [time-nuts] Tight PLL Tester



It is essential to understand exactly how this system works in theory.
No amount of hand waving or protestations will make its problems go 
away if you use inappropriate signal processing methods.


The tight PLL (or any other PLL) forces the VCO (VCOXO int this case) 
to servo the fluctuations in the phase difference between the test 
oscillator and the VCO to zero within the PLL bandwidth.


To recover the phase fluctuations (assuming linearity of the VCO 
response to its voltage control input) the EFC voltage has to be 
integrated.
Leaving aside the problems of saturation with most (but not all) 
integrators, the phase fluctuations at the output of the VCO can be 
recovered (to within a scale factor) by sampling the integrator 
output to produce a set of synthesized phase samples. Alternatively 
one can calculate the first differences of the periodic sequence of 
phase samples to produce a series of scaled frequency averages.


In practice integrator saturation can be avoided by one of the 
following methods:


1) Using a precision voltage to frequency converter and a counter to 
form the integrator.

This is how NIST used to do it.
The VFC110 from TI appears suitable.
Avoid using a synchronous VFC (eg AD652) as they suffer from 
injection locking effects:

http://www.analog.com/static/imported-files/tutorials/MT-028.pdf
However if one samples the VFC integrator output at the end of each 
integration the effect of injection locking can be corrected for.

DVMs like the HP/Agilent 34401A use a variation of this technique.
Another thing to be aware of is that a DVM may have a built in RC low 
pass filter between its input terminals and its ADC.
The effect of this may be significant if the averaging time 
(integration period) is too short.


2) Use an integrating DVM to sample the EFC voltage.
The DVM samples are equivalent (to within a scale factor) to a set of 
frequency average samples.
However most (but not all) DVMs have a finite deadtime between 
successive integrations, where the internal integrator is rundown for 
example.
If one uses an integrating DVM with finite deadtime then the 
calculated values of ADEV should be corrected using the bias 
functions tabulated in NBS special publication 140 and elsewhere:

http://digicoll.manoa.hawaii.edu/techreports/PDF/NBS140.pdf

3) The PLL has a finite bandwidth so one can sample it at a 
sufficiently high rate (> 2X PLL bandwidth as the PLL isnt a 
brickwall filter) and calculate the required frequency averages from 
the sampled data. Unless a very high oversampling rate is used merely 
averaging the values of a fixed number of samples will be 
insufficiently accurate. Attempts to use an arbitrary low pass filter 
to average the samples will bias the results. The averaging filter 
must have a frequency response that is very close to the sinc 
response of an integrator with an integration period equal to the 
sample interval.
However this method is the most expensive as a high resolution ADC 
capable of relatively high sampling rates (10x the PLL bandwidth??) 
is required.


It is also essential to have sufficient isolation between the unit 
under test and the VCO to avoid significant mutual injection locking 
effects.
To a first approximation such injection locking affects the PLL 
parameters so that the PLL loop parameters need to be measured whilst 
the PLL is closed when isolation is insufficient.


If one uses one of the Minicircuits phase detectors rather than an 
arbitrary mixer then the isolation between the phase detector inputs 
is much higher (at low frequencies at least) than that for most 
mixers. Depending on the reverse isolation of

Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier

[Bruce Griffiths]

Yes implementing an exact copy without using a hybrid would be difficult.
However for 10MHz use, its probably not too difficult since that 
isolation amplifier is intended for a 100MHz signal and the requirement 
is for 10MHz operation.


If the transistor ft's are reduced by a factor of 10 or so it shouldn't 
be too much of a problem.

At 10MHz 2N3906 and 2N3904 transistors should suffice.

Bruce

Bob Camp wrote:

Hi

Implementing that circuit without using a hybrid would be a bit of a challenge.

Bob


On Feb 10, 2010, at 11:30 PM, Bruce Griffiths wrote:

   

Clay

Circuit schematic for a more recent JPL isolation amplifier design is attached.

Bruce

life speed wrote:
 

Avoiding transformers and inductors will make it virtually impossible to
achieve very low phase noise as the dc gain from say the base of any
transistor in the chain to the output will degrade the flicker phase
noise. Using transformers or using an inductor to shunt any collector
resistors reduces the flicker phase modulation to low levels.

JPL in the past has built capacitively coupled complementary symmetry
isolation amplifiers that avoid transformers but suffer from dc loop
gains of around 3 or so.

Using complementary symmetry can be a good way of keeping the dc current
down.

How much reverse isolation do you need?
How low does the phase noise floor need to be?
What about flicker phase noise, how low does that need to be?

Bruce

Right, what do I really need? I only have a really good 10 MHz OCXO crystal 
oscillator to distribute, so about -120 dBc at 10 Hz, -140 dBc/Hz at 100 Hz, - 
150 dBc/Hz at 1KHz, and -155 dBc/Hz noise floor.  No maser or cesium clock, 
living in the world of practical realities here.  Of course I would like to be 
3 - 6 dB better than the OCXO numbers.

Reverse isolation is my primary interest in the distribution amplifier 
approach, although the OCXO is good enough that a sloppy approach could 
contaminate the phase noise also.  I would like to accomplish at least 100 dB 
reverse isolation at frequencies below 20 MHz, but more is better in this case. 
 The 10 MHz is running all over a noisy aircraft, to potentially noisy 
receivers.

In reading up on the subject, I have come to understand that DC gain is the 
bane of close-in phase noise.  Given that flicker noise is such a headache for 
we frequency synthesizer designers, I guess this should come as no surprise.

Clay (AKA Lifespeed)




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Re: [time-nuts] Vibrations

[Raj]

Hi Stan,

I will explore this issue tomorrow, 20:00 here now. I was just going 
through a boyish thrill of fiddling with a Sony ICF-S10Mk2 and the amazing 
numbers of AM stations it picks up.. maybe it can be used at 60Khz.. for 10$ it 
worth it !!

How about keeping he Rb cover open ? I measured the lamp housing at 
110C and the rest at 60-70C. I dont intend to keep my units on all the time but 
I dont feel good about anything running hot! Maybe a small slow fan might help. 
I am fascinated by the blinking light inside at 1Hz and thats not available on 
the DB connector. I did add a reverse polarity protection by soldering a 2A 
diode between the center pin of the regulator and the +15 pin. I wish I can 
find a schematic and what else I can do with this gizmo!

73 Raj vu2zap

At 11-02-10, you wrote:
>Hello Raj,
>
>I integrated a Symmetricom model X72 Rb  into a Down East Microwave (DEMI) 
>chassis
>and it does experience microphonics. This was for a friends 10 GHz transceiver 
>used for portable work.
>So getting rid of the microphonics is important.
>Somehow I need to mechanically isolate the Rb from the chassis and still have 
>a heat sink.
>
>Stan, W1LE
>
>
>Raj wrote:
>>While comparing a Rb FE5680 frequency reference with another RB or GPSDO I 
>>find that tapping a Rb unit causes a sudden shift in the scope display 
>>meaning the frequency has slightly shifted momentarily and locks back 
>>steadily with a phase shift. This does not happen in another Rb FE5680.
>>
>>Would something be loose I wonder ? Any experiences?
>>
>>Cheers
>>
>>  
>
>
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>and follow the instructions there.

-- 
Raj, VU2ZAP
Bangalore, India. 


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Re: [time-nuts] Vibrations

[Stan, W1LE]

Hello Raj,

I integrated a Symmetricom model X72 Rb  into a Down East Microwave 
(DEMI) chassis
and it does experience microphonics. This was for a friends 10 GHz 
transceiver used for portable work.

So getting rid of the microphonics is important.
Somehow I need to mechanically isolate the Rb from the chassis and still 
have a heat sink.


Stan, W1LE


Raj wrote:

While comparing a Rb FE5680 frequency reference with another RB or GPSDO I find 
that tapping a Rb unit causes a sudden shift in the scope display meaning the 
frequency has slightly shifted momentarily and locks back steadily with a phase 
shift. This does not happen in another Rb FE5680.

Would something be loose I wonder ? Any experiences?

Cheers

  



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Re: [time-nuts] TBolt self survey

[Arnold Tibus]

Raj,

don't trust the height asl from (older) maps. There are so many different local 
sea level definitions around the world, which are not yet WGS84/ EGM96 
conform (WGS 1972 would be good as well)! 
I had similar problems comparing swiss and and german map values 
with the Shuttle Radar Topography Mission (SRTM 2000) values.

On http://www.colorado.edu/geography/gcraft/notes/datum/edlist.html 
I found interesting informations about the differences (tables).
Eg. India used the Everest ellipsoid (India 1956, for Inda and Nepal) 
where the differences to WGS84 are (d = delta): 
dX= 295, dY= 736, dZ= 257 meters, what might explain already some 
discrepancies. Attention to the horizontal displacements as well!

So nothing to worry, as already correctly explained by several other 
fellow Time Nuts, use the elevation found with the auto position function
in LH, this is the right GPS value for this purpose ( perhaps moving around 
up to 1 m or similar), which is of no concern I think. Mark Sims and 
John Miles did really a good job! 
(The self survey function in Thunderbolt Monitor does give a similar 
result with somewhat less resolution/ precision.)

Searching around I found another page with good informations for 
beginners: 'Datum of a GPS Receiver'
http://homepages.slingshot.co.nz/~geoff36/datum.htm

The therein mentioned calculator 
'Geoid Height Calculator NGA EGM96', url:
http://earth-info.nga.mil/GandG/wgs84/gravitymod/egm96/intpt.html
does give the same results...

enjoy the time

Arnold, DK2WT

On Thu, 11 Feb 2010 06:26:42 +0530, Raj wrote:

>My GPS reading is about 850 +- 1
>The "official" height ASL = 920

>This calculator gives me ~936

>So what should I use ?

>Thanks
>Raj

>At 10-02-10, you wrote:
>>Try to find your correct height here:
>>http://sps.unavco.org/geoid/
>>
>>If you put in your correct coordinates without any input for the elevation, 
>>you will get theorthometric height, which is the GPS ellipsoidal height 
>>minus geoid height. That is the amount you need to subtract from 
>>your Thunderbolt GPS height in order to get your local geodetic height 
>>(WGS84). 
>>
>>Filling in the Thunderbolt GPS height you will get the 'real' height 
>>for your antenna...
>>
>>Some explanations:
>>http://www.unavco.org/edu_outreach/tutorial/geoidcorr.html
>>
>>On my location I have always to reduce the GPS height by 46.623 m.
>>Good luck with TB and the Lady Heather,
>>
>>Arnold, DK2WT
>>
>>
>>
>>On Wed, 10 Feb 2010 14:20:36 +0100, EB4APL wrote:
>>
>>>Raj,
>>
>>>Ask them about ellipsoidal and orthometric heights, or the local 
>>>difference between the geoid and the WGS84 ellipsoid.  Also your long 
>>>and lat may change depending of the Geodetic Reference System chosen.  
>>>By default GPS receivers use WGS84 and ellipsoidal height.
>>
>>>73 de Ignacio, EB4APL
>>
>>>--
>>>Raj wrote:
 Thanks Stan,

 That worked, I did not see the  control menu properly!

 I set the co-ords to 0 0 0 and started survey. The Tbolt found the Lat 
 long withing a few minutes and only the altitude is about 70 M below the 
 official figure for my place. I live a a few hundred meters from Bangalore 
 center and I am at the same height.. must check with the survey dept. down 
 the road!!

 73
 Raj vu2zap


 At 10-02-10, you wrote:
   
> Hello Raj,
>
> using the trimble T'boltMON software:
>
> Start with a "factory reset" in the control menu
>
> follow previous guidance of enabling the "save position" capability,
>
> Then to restart a survey
>
> go to the control menu (on left),
> select "restart self survey"
>
>
> In lady Heather V3.0 beta of 21 Jan '10 ( available from the KE5FX 
> website):
> keyboard enter a "s", for survey
> then "s" for a standard survey,
> use the default of 2000 samples,  for a start
> Then "enter"
>
> Time for me to go to bed.
>
> Please advise of progress.
>
> Stan,  W1LE
>
>
> Raj wrote:
> 
>> Thanks Stan,
>>
>> I did that but I can't where to "start the survey" in TBolt. In lady 
>> heather there
>> is a key for it.
>>
>> At 09-02-10, you wrote:
>>  
>>   
>>> Hello Raj,
>>>
>>> that location is probably where the coordinates were last saved. or 
>>> there abouts
>>> Or those are the default factory settings from memory,
>>> and new coordinates were never saved.
>>>
>>> redo the survey and save the results.
>>>
>>> using T'boltMON V2.60 on a PC:
>>>
>>> go to menu "set up",  then "self survey",   click on the "save position 
>>> flag"
>>> then "set survey",
>>> then "save segment",
>>> then "close"
>>>
>>> redo the survey and afterwards verify the local coordinates were saved.
>>>
>>> Lady Heather (software) c

Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier

[Bob Camp]

Hi

Implementing that circuit without using a hybrid would be a bit of a challenge. 

Bob


On Feb 10, 2010, at 11:30 PM, Bruce Griffiths wrote:

> Clay
> 
> Circuit schematic for a more recent JPL isolation amplifier design is 
> attached.
> 
> Bruce
> 
> life speed wrote:
>> Avoiding transformers and inductors will make it virtually impossible to
>> achieve very low phase noise as the dc gain from say the base of any
>> transistor in the chain to the output will degrade the flicker phase
>> noise. Using transformers or using an inductor to shunt any collector
>> resistors reduces the flicker phase modulation to low levels.
>> 
>> JPL in the past has built capacitively coupled complementary symmetry
>> isolation amplifiers that avoid transformers but suffer from dc loop
>> gains of around 3 or so.
>> 
>> Using complementary symmetry can be a good way of keeping the dc current
>> down.
>> 
>> How much reverse isolation do you need?
>> How low does the phase noise floor need to be?
>> What about flicker phase noise, how low does that need to be?
>> 
>> Bruce
>> 
>> Right, what do I really need? I only have a really good 10 MHz OCXO crystal 
>> oscillator to distribute, so about -120 dBc at 10 Hz, -140 dBc/Hz at 100 Hz, 
>> - 150 dBc/Hz at 1KHz, and -155 dBc/Hz noise floor.  No maser or cesium 
>> clock, living in the world of practical realities here.  Of course I would 
>> like to be 3 - 6 dB better than the OCXO numbers.
>> 
>> Reverse isolation is my primary interest in the distribution amplifier 
>> approach, although the OCXO is good enough that a sloppy approach could 
>> contaminate the phase noise also.  I would like to accomplish at least 100 
>> dB reverse isolation at frequencies below 20 MHz, but more is better in this 
>> case.  The 10 MHz is running all over a noisy aircraft, to potentially noisy 
>> receivers.
>> 
>> In reading up on the subject, I have come to understand that DC gain is the 
>> bane of close-in phase noise.  Given that flicker noise is such a headache 
>> for we frequency synthesizer designers, I guess this should come as no 
>> surprise.
>> 
>> Clay (AKA Lifespeed)
>> 
>> 
>> 
>> 
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>> and follow the instructions there.
>> 
>>   
> 
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Re: [time-nuts] Vibrations

[Bob Camp]

Hi

Sounds like something loose. Could be something simple. 

Bob


On Feb 11, 2010, at 6:45 AM, Raj wrote:

> 
> While comparing a Rb FE5680 frequency reference with another RB or GPSDO I 
> find that tapping a Rb unit causes a sudden shift in the scope display 
> meaning the frequency has slightly shifted momentarily and locks back 
> steadily with a phase shift. This does not happen in another Rb FE5680.
> 
> Would something be loose I wonder ? Any experiences?
> 
> Cheers
> 
> -- 
> Raj, VU2ZAP
> Bangalore, India. 
> 
> 
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> 


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[time-nuts] Vibrations

[Raj]

While comparing a Rb FE5680 frequency reference with another RB or GPSDO I find 
that tapping a Rb unit causes a sudden shift in the scope display meaning the 
frequency has slightly shifted momentarily and locks back steadily with a phase 
shift. This does not happen in another Rb FE5680.

Would something be loose I wonder ? Any experiences?

Cheers

-- 
Raj, VU2ZAP
Bangalore, India. 


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Re: [time-nuts] injection locking

[Joop]

Kit VK2LL wrote:
>Theory suggests that lock sensitivity could be improved in this instance
>if the 10MHz duty cycle was changed from 1:1 to 6 or 7:1, the idea being
>that the narrower pulse should be <0.5 period of the frequency of the
>oscillator to be locked.  

Ah, that triggers some thought.
I know from (ARDF) experiments that sub-critical oscillators can be used
as very narrow filters. Just as the Q-multiplier circuits.
Even at the point where there is enough gain and oscillation starts, I
found the oscillator tracks and locks onto the incoming frequency.

Thinking this way of injection locking, it makes sense to increase
harmonic content of the injected signal. In essence it will then not lock
onto the fundamental frequency, but on some harmonic part of it. In my
SPICE investigation I injected a clean sine signal. Perhaps I should try
with some clipped or digital signal.
That might also explain why James G1PVZ was so successful. The bottom
2N918 might cause a lot of harmonics if driven into the collector cut-off.

Joop PE1CQP


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Re: [time-nuts] Tight PLL Tester

[WarrenS]

Bruce

Thanks for your response, as always you've give me plenty to think about.

Bruce said:  It is essential to understand exactly how this system works in 
theory.

Turns out to be too true.
After re-reading your last post several times,
I now finally understand what you are saying and why you are saying it.
It is because YOU do not yet understand how this method works.
I find that so unbelievable, that I had not considered that possibility.

Starting at the following line and pretty much everything after that, 
although accurate statements,

THEY DO NOT APPLY to this method.

To recover the phase fluctuations the EFC voltage has to be integrated.

...

ws

***
- Original Message - 
From: "Bruce Griffiths" 
To: "Discussion of precise time and frequency measurement" 


Sent: Wednesday, February 10, 2010 2:32 PM
Subject: Re: [time-nuts] Tight PLL Tester



It is essential to understand exactly how this system works in theory.
No amount of hand waving or protestations will make its problems go away 
if you use inappropriate signal processing methods.


The tight PLL (or any other PLL) forces the VCO (VCOXO int this case) to 
servo the fluctuations in the phase difference between the test oscillator 
and the VCO to zero within the PLL bandwidth.


To recover the phase fluctuations (assuming linearity of the VCO response 
to its voltage control input) the EFC voltage has to be integrated.
Leaving aside the problems of saturation with most (but not all) 
integrators, the phase fluctuations at the output of the VCO can be 
recovered (to within a scale factor) by sampling the integrator output to 
produce a set of synthesized phase samples. Alternatively one can 
calculate the first differences of the periodic sequence of phase samples 
to produce a series of scaled frequency averages.


In practice integrator saturation can be avoided by one of the following 
methods:


1) Using a precision voltage to frequency converter and a counter to form 
the integrator.

This is how NIST used to do it.
The VFC110 from TI appears suitable.
Avoid using a synchronous VFC (eg AD652) as they suffer from injection 
locking effects:

http://www.analog.com/static/imported-files/tutorials/MT-028.pdf
However if one samples the VFC integrator output at the end of each 
integration the effect of injection locking can be corrected for.

DVMs like the HP/Agilent 34401A use a variation of this technique.
Another thing to be aware of is that a DVM may have a built in RC low pass 
filter between its input terminals and its ADC.
The effect of this may be significant if the averaging time (integration 
period) is too short.


2) Use an integrating DVM to sample the EFC voltage.
The DVM samples are equivalent (to within a scale factor) to a set of 
frequency average samples.
However most (but not all) DVMs have a finite deadtime between successive 
integrations, where the internal integrator is rundown for example.
If one uses an integrating DVM with finite deadtime then the calculated 
values of ADEV should be corrected using the bias functions tabulated in 
NBS special publication 140 and elsewhere:

http://digicoll.manoa.hawaii.edu/techreports/PDF/NBS140.pdf

3) The PLL has a finite bandwidth so one can sample it at a sufficiently 
high rate (> 2X PLL bandwidth as the PLL isnt a brickwall filter) and 
calculate the required frequency averages from the sampled data. Unless a 
very high oversampling rate is used merely averaging the values of a fixed 
number of samples will be insufficiently accurate. Attempts to use an 
arbitrary low pass filter to average the samples will bias the results. 
The averaging filter must have a frequency response that is very close to 
the sinc response of an integrator with an integration period equal to the 
sample interval.
However this method is the most expensive as a high resolution ADC capable 
of relatively high sampling rates (10x the PLL bandwidth??) is required.


It is also essential to have sufficient isolation between the unit under 
test and the VCO to avoid significant mutual injection locking effects.
To a first approximation such injection locking affects the PLL parameters 
so that the PLL loop parameters need to be measured whilst the PLL is 
closed when isolation is insufficient.


If one uses one of the Minicircuits phase detectors rather than an 
arbitrary mixer then the isolation between the phase detector inputs is 
much higher (at low frequencies at least) than that for most mixers. 
Depending on the reverse isolation of the output buffers of the 
oscillators being compared this isolation may be sufficient to avoid an 
appreciable change in the PLL parameters. If the isolation is insufficient 
one then needs to use a suitable isolation amplifier between the the 
output of each oscillator and the phase detector.
The phase noise of the isolation amplifier should be lower than that of 
the reference VCO (VCOCXO in this case).
Suitable isolation amplifiers are readil

Re: [time-nuts] Injection locking

[Rex]

Kit Scally wrote:

...

On a related topic, I found some while ago - and promptly lost - a
graph/chart showing harmonic level variations with varying duty-cycle of
an input waveform.  This was to some degree a graphical representation
of the Wenzel document referenced by Bruce recently.  Has anyone got a
link to this document please ?


Kit
VK2LL 

  


Is this the Wenzel document you are referring to?
http://www.wenzel.com/pdffiles1/pdfs/choose.pdf  ("Choosing a Frequency 
Multiplier's Waveform")


Figure 2 in that doc is a graph of Harmonic Amplitudes vs. pulse width. 
Is that what you were looking for? There are a couple of bar charts too, 
for specific duty cycles.





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Re: [time-nuts] Tight PLL Tester

[WarrenS]

Tom

Things will turn out much better to do it the other way around.
When I find out who is going to build/test it,
I'll do something special for them and their tool box.

ws

**

- Original Message - 
From: "Tom Van Baak" 




If there are any Nuts out there interested in helping to make available 
to other Freq-Nuts a SIMPLE tester that I have found to be a VERY useful 
low cost tool,


Warren,

Yes, I think it's a good idea for a couple of people to try to
duplicate your results; either to validate the resolution and
features that you're claiming, or to locate or quantify the
limitations in your implementation. Either way it will be a
learning experience for you, and for the group.

To that end, would you be able this week to write a quick
word document or readme or web page with photo(s) of
your setup, schematic, parts list, specific make/model of
the equipment that you're using, etc. Since you say it is
a simple setup, I suspect a number of us would then be
able to dig in our parts bin and mimic your prototype
as close as possible and then objectively measure how
it works compared to other phase noise measurement
systems.

/tvb




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[time-nuts] distribution amplifier specifications

[Clive Green]

Our standard A5 distribution amplifier, > 12 years old has >90dB adjacent
out to out i...@10mhz, -130dB non adj iso @5MHz & out to in >-11...@10mhz. in
to in (crosstalk is >9...@5mhz) Spur >-110, BB noise -148dBm/Hz, delay match
<2ns (or 300ps within a group of 4 ops), phase/temp of 10ps/C, open/short
any o/p gives phase change 0.5ps, however, most important specs are PN @1Hz
-140dBc/Hz, 10Hz -150dBc/Hz, >100Hz -165dBc/Hz and AVAR mailto:n...@quartzlock.com> cgr...@quartzlock.com ü:
 www.quartzlock.com

Skype: clive.green.skype Messenger: cgr...@quartzlock.com

Registered office: Gothic, Plymouth Road, Totnes, Devon. TQ9 5LH England
Registered in England

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Re: [time-nuts] Symmetricom Announces Commercial Time-Scale System

[John Miles]

> -Original Message-
> From: time-nuts-boun...@febo.com [mailto:time-nuts-boun...@febo.com]on
> Behalf Of Poul-Henning Kamp
> Sent: Thursday, February 11, 2010 12:29 AM
> To: Discussion of precise time and frequency measurement
> Subject: Re: [time-nuts] Symmetricom Announces Commercial Time-Scale
> System
>
>
> In message <4b73b011.5070...@murgatroid.com>, Christopher Hoover writes:
> >fyi.  -ch.
> >
> >
> >  Symmetricom Announces Commercial Time-Scale System
>
> I'm pretty sure this is just timing.com's product that gets
> a new outing...
>

It does look very similar to what the LORAN stations use (having just seen a
rack of gear just like it during a visit).

-- john, KE5FX


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Re: [time-nuts] Advice on 10 MHz isolation/distribution amplifier

[Bruce Griffiths]

Clay

If the first stage has a voltage gain of 2x then the total dc current 
can be reduced as the output stages no longer need to drive the feedback 
100 ohm resistor connected to ground at RF. The attached circuit 
schematic also includes faster input transistors in each 3 transistor 
feedback circuit to improve stability and increase the reverse isolation 
slightly.


Bruce

Bruce Griffiths wrote:

Clay

You could try something like the attached circuit schematic.
Austron used buffer amplifiers like this albeit without the 
complementary symmetry output stage.

There are no transformers and the dc gain is low.
Simulated reverse isolation at 10MHz is around 120dB.
Simulated crosstalk between the 2 outputs is around -100dB at 10MHz.
The transistor models used usually predict reverse isolation 
reasonably accurately at 10MHz.
The phase noise floor should be around -170dBc/Hz or less at 100kHz 
offset.


V1 is the input signal.

The 50 ohm sources  V6, V7 shown at the outputs are used for 
simulation purposes (reverse isolation and crosstalk).


Off course, more elaborate power supply decoupling will be necessary 
to avoid degrading reverse isolation and crosstalk.


If you are really desperate to reduce the dc current the output 
transistors could be operated in class B.

However the distortion will increase a little.

Bruce

life speed wrote:

Avoiding transformers and inductors will make it virtually impossible to
achieve very low phase noise as the dc gain from say the base of any
transistor in the chain to the output will degrade the flicker phase
noise. Using transformers or using an inductor to shunt any collector
resistors reduces the flicker phase modulation to low levels.

JPL in the past has built capacitively coupled complementary symmetry
isolation amplifiers that avoid transformers but suffer from dc loop
gains of around 3 or so.

Using complementary symmetry can be a good way of keeping the dc current
down.

How much reverse isolation do you need?
How low does the phase noise floor need to be?
What about flicker phase noise, how low does that need to be?

Bruce

Right, what do I really need? I only have a really good 10 MHz OCXO 
crystal oscillator to distribute, so about -120 dBc at 10 Hz, -140 
dBc/Hz at 100 Hz, - 150 dBc/Hz at 1KHz, and -155 dBc/Hz noise floor.  
No maser or cesium clock, living in the world of practical realities 
here.  Of course I would like to be 3 - 6 dB better than the OCXO 
numbers.


Reverse isolation is my primary interest in the distribution 
amplifier approach, although the OCXO is good enough that a sloppy 
approach could contaminate the phase noise also.  I would like to 
accomplish at least 100 dB reverse isolation at frequencies below 20 
MHz, but more is better in this case.  The 10 MHz is running all over 
a noisy aircraft, to potentially noisy receivers.


In reading up on the subject, I have come to understand that DC gain 
is the bane of close-in phase noise.  Given that flicker noise is 
such a headache for we frequency synthesizer designers, I guess this 
should come as no surprise.


Clay (AKA Lifespeed)


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Re: [time-nuts] Symmetricom Announces Commercial Time-Scale System

[Poul-Henning Kamp]

In message <4b73b011.5070...@murgatroid.com>, Christopher Hoover writes:
>fyi.  -ch.
>
>
>  Symmetricom Announces Commercial Time-Scale System

I'm pretty sure this is just timing.com's product that gets
a new outing...

Poul-Henning


-- 
Poul-Henning Kamp   | UNIX since Zilog Zeus 3.20
p...@freebsd.org | TCP/IP since RFC 956
FreeBSD committer   | BSD since 4.3-tahoe
Never attribute to malice what can adequately be explained by incompetence.

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